Delay lock clock synthesizer and method thereof

ABSTRACT

A delay lock clock synthesizer comprises: an adjustable delay circuit for receiving an input clock and for generating an output clock having a phase offset controlled by a control signal; a phase detector for detecting a phase difference between the input clock and the output clock and for generating a phase error signal representing the phase difference; a summing circuit for summing the phase error signal and a phase offset signal into a modified phase error signal; and a filter for filtering the modified phase error signal to generate the control signal to control the adjustable delay circuit.

REFERENCE TO RELATED APPLICATIONS

This application claims priority of U.S. Provisional Patent ApplicationSer. No. 60/745,188, filed on Apr. 20, 2006, and is related to thefollowing copending application, owned by the assignee of thisinvention:

-   -   1) Lin et al, Ser. No. 11/517,415, for “VARIABLE DELAY CLOCK        CIRCUIT AND METHOD THEREOF”.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a method and apparatus for generating avariable delay clock and in particular to a system of controlling thedelay of a clock with high resolution in the delay.

2. Description of Related Art

DLL (delay lock loop) is well known in prior art for clock generation.FIG. 1 depicts a functional block diagram of a typical N-stage DLL 100,which comprises: a VCDL (voltage-controlled delay line) 110, a PD (phasedetector) 120, and a LF (loop filter) 130. VCDL 110 further comprises Nvariable delay cells 111_1, 111_2, and so on. VCDL 110 receives an inputclock CLK_IN and a control voltage Vc from LF 130, and generates Noutput clocks CLK_1, CLK_2, and so on. CLK_1 is the output of the 1^(st)variable delay cell 111_1, CLK_2 is the output of the 2^(nd) variabledelay cell 111_2, and so on. All N delay cells (111_1, 111_2, and so on)are constructed from substantially the same circuit; therefore they allcause substantially the same amount of delay to their respective inputs.The phase of the output clock CLK_N from the last variable delay cell111_N is compared with the phase of the input clock CLK_IN by the PD120, which generates a phase error signal PE indicative of the phaserelationship between the input clock CLK_IN and the output clock CLK_N.The phase error signal PE generated by PD 120 is filtered by the LF 130,resulting in the control voltage Vc to control the delay for each of theN delay cells of VCDL 110. In steady state, a steady control voltage Vcis established so that the output clock CLK_N is aligned with the inputclock CLK_IN; the phase error signal PE is virtually zero, indicating nofurther change to the control voltage Vc is needed. Let the period ofthe input clock CLK_IN be T. In steady state, each delay cell (111_1,111_2, and so on) must cause a delay of TIN so that CLK_N can be alignedwith CLK_IN. In many applications, a phase inversion operation (notshown in FIG. 1) is performed at the output of the last delay cell togenerate an additional 180-degree phase shift (or equivalent T/2 delay).In this case, each delay cell (111_1, 111_2, and so on) causes a delayof T/(2N) in steady state.

A clock multiplexer is often used along with a DLL to generate a clockof a variable phase (or delay). A clock generation system 200constructed using a N-stage DLL 100 and a clock multiplexer 220 isillustrated in FIG. 2. N-stage DLL 100 receives an input clock CLK_INand generates N output clocks CLK_1, CLK_2, and so on, in a mannerillustrated in FIG. 1. Clock multiplexer 220 receives those N outputclocks from N-stage DLL 100 along with a control signal PHASE_SELECT,and generates CLK_OUT as the output clock of the clock generation system200. The output clock CLK_OUT is selected among the N output clocksCLK_1, CLK_2, and so on, based on the PHASE_SELECT signal.

Although prior art clock generation system 200 can generate a clock witha desired phase (or delay), there are two problems. First, a clockmultiplexer circuit is needed. A high frequency clock multiplexer ishard to implement in an integrated circuits, especially when the numberof inputs is high. Second, the resolution of the delay depends on thenumber of stages of delay buffers. In general, a N-stage DLL (with anaforementioned phase inversion at the output of the last delay cell)provides a resolution of 180/N degrees in phase delay. To achieve a10-degrees resolution of phase delay, for instance, it takes an 18-stageDLL. Therefore, it is impractical to use DLL to generate a variabledelay clock with high resolution in the phase delay.

What is needed is a clock generation system that offers a highresolution in clock phase yet does not require a high complexity phasemultiplexer.

BRIEF SUMMARY OF THIS INVENTION

In an embodiment, an apparatus is disclosed, the apparatus comprising:an adjustable delay circuit for receiving an input clock and forgenerating an output clock having a phase offset controlled by a controlsignal; a phase detector for detecting a phase difference between theinput clock and the output clock and for generating a phase error signalrepresenting the phase difference; a summing circuit for summing thephase error signal and a phase offset signal into a modified phase errorsignal; and a filter for filtering the modified phase error signal togenerate the control signal to control the adjustable delay circuit.

In an embodiment, a method for generating an output clock is disclosed,the method comprising: receiving an input clock; generating the outputclock by delaying the input clock by an amount of delay controlled by acontrol signal; detecting a phase difference between the input clock andthe output clock to generate a phase error signal; summing the phaseerror signal and an offset signal into a modified phase error signal;and filtering the modified phase error signal to generate the controlsignal.

BRIEF DESCRIPTION OF THE DRAWINGS

The subject matter regarded as the invention is particularly pointed outand distinctly claimed in the concluding portion of the specification.The invention, both as to device and method of operation, together withfeatures and advantages thereof may best be understood by reference tothe following detailed description with the accompanying drawings inwhich:

FIG. 1 depicts a functional block diagram of a typical N-stage delaylock loop (DLL);

FIG. 2 depicts a functional block diagram of a typical clock generationsystem;

FIG. 3 illustrates an embodiment of a delay clock synthesizer (DLCS)according to the present invention;

FIG. 4 depicts an exemplary embodiment of the phase detector (PD) ofFIG. 3;

FIG. 5 illustrates an exemplary embodiment for generating the phaseoffset signal PO;

FIG. 6 shows a timing diagram for this instance under various PHA_OSvalues;

FIG. 7 shows an exemplary variable delay clock synthesizer according tothe present invention;

FIG. 8 shows an exemplary timing diagram for a case where STATE=0 andPOX=I/4;

FIG. 9 depicts an exemplary embodiment of FSM according to the presentinvention; and

FIG. 10 depicts an exemplary embodiment of crossover detector accordingto the present invention.

DETAILED DESCRIPTION OF THIS INVENTION

The present invention relates to a method and apparatus for controllingthe phase delay of a clock with high resolution in the delay. While thespecifications described several example embodiments of the inventionconsidered best modes of practicing the invention, it should beunderstood that the invention can be implemented in many way and is notlimited to the particular examples described below or to the particularmanner in which any features of such examples are implemented.

A delay clock synthesizer (DLCS) in accordance with the presentinvention is illustrated in FIG. 3. In this embodiment, DLCS 300receives an input clock CLK_IN and a phase offset signal PO, andgenerates an output clock CLK_OUT, which has a phase offset relative tothe input clock CLK_IN, wherein the phase offset is controlled by the POsignal. DLCS 300 comprises a phase detector PD 310, a summing circuit320, a loop filter LF 330, and a voltage controlled delay line VCDL 340.The VCDL 340 receives the input clock CLK_IN and generates the outputclock CLK_OUT by delaying the input clock by an amount controlled by acontrol voltage Vc provided from the loop filter LF 330. The phasedetector PD 310 compares a phase of the input clock CLK_IN with a phaseof the output clock CLK_OUT and generates accordingly a phase errorsignal PE representing the phase difference between CLK_IN and CLK_OUT.The phase error signal PE is summed with the PO signal by the summingcircuit 320, resulting in a modified phase error signal PE′. Themodified phase error signal PE′ is filtered by the loop filter LF 330,resulting in the control voltage Vc. In a closed-loop manner, the phaseof CLK_OUT is adjusted to establish a certain relationship with thephase of CLK_IN. In steady state, the phase of CLK_OUT settles to acertain value relative to the phase of CLK_IN so that the phase errorsignal PE is virtually offset by the phase offset signal PO; as aresult, the modified phase error signal PE′ is virtually zero,indicating no further change to the phase of CLK_OUT is needed. In anembodiment, loop filter LF 330 comprises a capacitor.

In a preferred embodiment, both the phase error signal PE and the phaseoffset signal PO are current signals. In this case, both signals can bedirectly tied together to generate the modified phase error signal PE′without using an explicit summing circuit 320.

In a preferred embodiment, PD 310 is implemented as a linear phasedetector; every time a phase comparison is made, PD 310 generates apulse of a fixed magnitude but a variable width proportional to thephase difference between CLK_IN and CLK_OUT. The polarity of the pulseindicates the timing relationship between CLK_IN and CLK_OUT; forexample, the pulse is positive if CLK_OUT is earlier than CLK_IN, and isnegative otherwise. In a preferred embodiment, the pulse is implementedas an electrical current pulse.

An exemplary embodiment for implementing PD 310 of FIG. 3 is depicted inFIG. 4. Here, PD 400 comprises a phase-frequency detector PFD 410 (whichis an example of a linear phase detector) and a charge pump circuit CP420. PFD 410 receives two clock signals: CLK_IN (which is the inputclock of DLCS 300 of FIG. 3) and CLK_OUT (which is the output clock ofDLCS 300 of FIG. 3), and generates accordingly two logical signals: UPand DN. PFD 410 comprises two data flip-flops (DFF) 412 and 414, and anAND gate 416. Each DFF has four terminals: input terminal D, clocktriggering terminal, reset terminal R, and output terminal Q. Theprinciple of PFD is well known in prior art and thus not explained indetail here. The charge pump circuit CP 420 comprises a current source422 of magnitude I, a first switch 424 controlled by the UP signal, asecond switch 426 controlled by the DN signal, and a current sink 428 ofmagnitude I. The principle of charge pump circuit is also well known inprior art and thus not explained in detail here. When the timing ofCLK_OUT is earlier than the timing of CLK_IN by an amount τ, a positivecurrent pulse of magnitude I and width τ is generated in the phase errorsignal PE; when the timing of CLK_IN is earlier than the timing ofCLK_OUT by an amount τ, a negative current pulse of magnitude I andwidth τ is generated in the phase error signal PE.

The phase offset signal PO is preferably generated by a DAC(digital-to-analog converter). An exemplary embodiment for generatingthe phase offset signal PO using a DAC 500 is illustrated in FIG. 5. Inthis embodiment, the phase offset is represented by an integer PHA_OS,where −K≦PHA_OS≦K and K is a positive integer. An encoder convertsPHA_OS into K ternary codes P₁, P₂, and so on. Each ternary code hasthree possible values, say −1, 0, and 1. The encoder works in a mannersuch that the sum of all K ternary codes equals PHA_OS. Each ternarycode is received and converted into an analog signal by a ternary DAC(digital-to-analog converter). For example, P₁ is received and convertedby DAC 520_1, P₂ is received and converted by DAC 520_2, and so on. Theoutputs from all ternary DAC are summed by a summing circuit 530,resulting in the phase offset signal PO. In a preferred embodiment, allternary DAC are current-mode digital-to-analog converters, and theiroutputs can be directly tied together to generate the phase offsetsignal PO without using an explicit summing circuit 530. Note that onemay also choose to use an alternative encoder to convert PHA_OS into aplurality of binary codes, each having two possible values (say −1and 1) without departing from the principle of the present invention.Or, one may also choose to use yet an alternative encoder to converterPHA_OS into a combination of binary and ternary codes without departingfrom the principle of the present invention.

Still refer to FIG. 5. In a preferred embodiment, each ternary DAC(520_1, 520_2, and so on) is implemented using a corresponding chargepump circuit similar to CP 420 of FIG. 4. Each of the K ternary codes(P₁, P₂, and so on) is represented by two logical signals (see UP and DNof FIG. 4): one to control a first switch (see 424 of FIG. 4) thatenables the corresponding charge pump to source a current, and the otherto control a second switch (see 426 of FIG. 4) that enables thecorresponding charge pump to sink a current. For instance, when theternary code is 1 (UP=1 and DN==0), the corresponding charge pumpsources an outgoing current; when the ternary code is −1 (UP==0 andDN==1), the corresponding charge pump sinks an incoming current; whenthe ternary code is 0 (UP==0 and DN==0), the corresponding charge pumpcircuit is effectively disabled. In an exemplary embodiment, the currentoutput from each ternary DAC implemented by a corresponding charge pumpis: (1) J when the ternary code is 1, (2) −J when the ternary code is−1, and (3) zero when the ternary code is 0. The resultant value of theoutput current representing the PO signal is thus PHA_OS·J. Now referback to FIG. 3. In steady state, the PE signal has to be offset by thePO signal, i.e. their time-averages (or time-integrals) must be the samebut of opposite signs. Let the timing difference between CLK_IN andCLK_CLOCK be τ, then we have the following relation in steady stateusing a principle of charge conservation:τ·I=PHA _(—) OS·J·Torτ=T·PHA _(—) OS·J/I.

Here, I is the current magnitude of the charge pump within the phasedetector (see FIG. 4), J is the charge pump current magnitude for eachof the ternary DAC from which the phase offset signal PO is generated,PHA_OS an integer controlling the generation of the phase offset signalPO, and T is the period of CLK_IN. The quantity T·PHA_OS·J/I is indeedthe phase offset signal PO of FIG. 3 under the embodiment of FIG. 5.

In this manner, a desired phase difference between CLK_IN and CLK_OUTcan be established by choosing a proper PHA_OS. For instance, let PHA_OSbe an integer between −4 and 4, inclusively. (That is, K=4 for theexample in FIG. 5.) Let J be I/8. Then, the timing difference betweenCLK_IN and CLK_OUT will be T·PHA_OS/8 in steady state. A timing diagramfor this instance under various PHA_OS values is shown in FIG. 6. Toachieve a high resolution in delay, one simply needs to choose a largeK.

Note that the phase offset between the input clock CLK_IN and the outputclock CLK_OUT using the embodiment disclosed thus far is bounded within[−T, T], since the phase difference between two clocks of the samefrequency, as detected by a phase detector, cannot exceed the clockperiod. Therefore, the quantity PHA_OS·J/I also needs to be boundedwithin [−1, 1] to ensure the steady state condition PE′=0 is met. As aresult, the phase offset caused by the DLCS 300 is also bounded within[−T, T].

In some applications, it is desirable to synthesize a clock with a phaseoffset exceeding one full clock cycle. For a phase lock loopapplication, in particular, the amount of phase offset should beunbounded. In this case, it is more convenient to specify an amount ofcycle-to-cycle phase change, rather than an amount of absolute phaseoffset. By way of example without loss of generality, one uses a ternarysignal PHA_CH to indicate an incremental phase change (from last clockcycle), instead of using the PHA_OS signal to indicate an absolute phaseoffset. The ternary signal PHA_CH has three possible values: 0, 1, and−1. PHA_CH=0 indicates no phase change (from last clock cycle); PHA_CH=1indicates a further phase delay; and PHA_CH=−1 indicates a further phaseadvance. The absolute phase offset is a cumulative sum of the PHA_CHsignal and is thus unbounded.

In an embodiment, a clock generation system using a dual VDCC (variabledelay clock circuit) architecture is used to generate a clock with anunbounded phase offset. A dual VDCC architecture comprises two VDCC; inany moment of operation, one of the two VDCC is in an active state,while the other is in a stand-by state. The VDCC currently in the activestate is used for generating a final output clock for the clockgeneration system, while the VDCC currently in the stand-by state isused for generating a stand-by clock for the clock generation system.Initially, the phase difference between the final output clock and thestand-by clock is 180 degrees. The phase of the final output clock canbe adjusted by controlling a phase offset signal for the active VDCC.When the phase of the final output clock is adjusted to an extent thatthe phase offset equals 180 degrees, we exchange the roles of the twoVDCC. That is, the currently stand-by VDCC takes over the role forgenerating the final output clock, while the other VDCC enters into astand-by state. Each time we make an exchange of the roles of the twoVDCC, we effectively extend the range of phase offset of the finaloutput clock by 180 degrees. In this manner, the phase offset of thefinal output clock is unbounded.

An exemplary variable delay clock synthesizer 700 for achievingunbounded phase offset using a dual DLCS (which is an example of VDCC)architecture is shown FIG. 7. Here, variable delay clock synthesizer 700comprises two delay lock clock synthesizers (DLCS) 300_0 and 300_1, amultiplexer 720, and a finite state machine (FSM) 710. Both DLCS 300_0and 300_1 are constructed from the same circuit as DLCS 300 of FIG. 3.DLCS 300_0 receives an input clock CLK_IN and a first phase offsetsignal P0, and generates a first output clock CLK_OUT0, which has aphase offset relative to the input clock CLK_IN, the offset beingdetermined by P0. DLCS 300_1 receives an inverted input clock CLK_INB(which is 180 degrees out of phase relative to the input clock CLK_IN)and a second phase offset signal PO1, and generates a second outputclock CLK_OUT1, which has a phase offset relative to the input clockCLK_INB, the phase offset being determined by PO1. Multiplexer 720receives the first output clock CLK_OUT0 from DLCS 300_0 and the secondoutput clock CLK_OUT1 from DLCS 300_1, and generates a final outputclock CLK_OUT based on a logical signal STATE. When STATE is 0, CLK_OUT0is selected for the final output clock; otherwise, CLK_OUT1 is selected.FSM 710 receives the output clock CLK_OUT0 from DLCS 300_0, the outputclock CLK_OUT1 from DLCS 300_1, and a phase change signal PHA_CH, andgenerates accordingly the first phase offset signal PO0 to control thephase offset for DLCS 300_0, the second phase offset signal PO1 tocontrol the phase offset for DLCS 300_1, and the logical signal STATE todetermine which DLCS is selected for generating the final output clock.

The underlying principle of operation for variable delay clocksynthesizer 700 is described as follows. By way of example without lossof generality, the phase change signal PHA_CH is a ternary signal withthree possible values: 0, 1, and −1. Whenever PHA_CH is non-zero, aphase advance or delay is commanded. Inside FSM 710, there is an up/downcounter storing a phase offset variable POX. If PHA_CH is 1, POX isincremented; if PHA_CH is −1, POX is decremented. The DLCS currentlyselected for generating the final output clock is said to be in anactive state, while the other DLCS is said to be in a “stand-by” state.For the active DLCS, the value of the phase offset variable POX isassigned as its corresponding phase offset signal. For the stand-byDLCS, a value of zero (0) is assigned as its corresponding phase offsetsignal. For instance, when STATE is 0, DLCS 300_0 is in an active stateand one assigns the value of POX to the first phase offset signal P0; inthe meanwhile, DLCS 300_1 is in a stand-by state and one assigns zero(0) to the second phase offset signal PO1. When STATE is 1, DLCS 300_1is in an active state and one assigns the value of POX to the secondphase offset signal PO1; in the meanwhile, DLCS 300_0 is in a stand-bystate and one assigns zero (0) to the first phase offset signal P0. EachDLCS circuit works in a closed-loop manner to settle into a conditionwhere its phase error signal is canceled by the corresponding phaseoffset signal. For instance, for a case where STATE is 0, PE0 willsettle to POX and PE1 will settle to zero; as a result, CLK_OUT0 willhave a phase offset (relative to CLK_IN) determined by POX, and CLK_OUT1will have the same phase as CLK_INB. In this manner, the phase of theoutput clock from the active DLCS is thus advanced or delayed due to theincrement or decrement of the phase offset variable POX, while thestand-by DLCS will generate an output clock having the same phase as itscorresponding input clock. An exemplary timing diagram for a case whereSTATE=0 and POX=I/4 is shown in FIG. 8; which shows CLK_OUT1 has a 180degrees (T/2) delay and CLK_OUT0 has a 90 degrees (T/4) delay, bothrelative to the input clock CLK_IN.

If the magnitude of the phase offset variable POX reaches I/2,accordingly the phase delay or advance for the active DLCS also reachesT/2. This condition, referred to as “crossover,” can be detected, forexample, by making a phase comparison between CLK_OUT0 and CLK_OUT1, asCLK_OUT0 and CLK_OUT1 will align with each other at the instant wherethe phase delay/advance for the active DLCS reaches T/2. In this case,FSM 710 toggles the logical signal STATE, and resets POX, P0, and PO1 tozero.

FIG. 9 depicts an exemplary embodiment for FSM 710. In this embodiment,FSM 710 comprises an accumulator ACC 910, a DAC (digital-to-currentconverter) 920, a crossover detector 930, a flip-flop 940, a logicalinverter 950, a first multiplexer 960, and a second multiplexer 970. ACC910, which is an up/down counter, receives the ternary signal PHA_CH,which signals ACC 910 to count up, count down, or stay unchanged. Theoutput of ACC 910 is an integer signal PHA_OS, which is converted intoan electrical signal POX, preferably implemented as an electricalcurrent signal, by DAC 920, which is preferably implemented using thecircuit DAC 500 shown in FIG. 5. Crossover detector 930 receivesCLK_OUT0 from DLCS 300_0 and CLK_OUT1 from DLCS 300_1 and generates alogical signal RESET, which is provided for resetting the counter forACC 910 and for triggering flip-flop 940. Crossover detector 930 detectsthe condition of the crossover of the two clocks, CLK_OUT0 and CLK_OUT1. Whenever a crossover condition is detected, the RESET signal isasserted to reset the counter value for ACC 910. At the same time, theoutput of flip-flop 940 is toggled upon the triggering of the RESETsignal due to the inverting feedback connection via inverter 950. Theoutput of flip-flop 940, i.e. the STATE signal, is used to determinewhich DLCS is selected for generating the final output clock. When STATEis 0, DLCS 300_0 is selected; in this case, POX is assigned to PO0 viamultiplexer 960, and PO1 is set to zero via multiplexer 970. When STATEis 1, DLCS 300_1 is selected; in this case, POX is assigned to PO1 viamultiplexer 970, and PO0 is set to zero via multiplexer 960.

FIG. 10 depicts an exemplary embodiment of crossover detector 930, whichcomprises a first flip-flop 1060, a second flip-flop 1030, a XOR gate1040, an AND gate 1050, an ABS (absolute value) operator 1080, and acomparator 1090. CLK_OUT1 is used to sample CLK_OUT0 using flip-flop1060, resulting in a logical signal S1, which is further sampled byflip-flop 1030, resulting in a logical signal S2. When crossover occurs,i.e. CLK_OUT0 is aligned with CLK_OUT1, S1 will be a logical inversionof S2. The logical signal XO, which is obtained by an XOR operation onS1 and S2 using the logical gate 1040, will be asserted. However, it isobvious to those of ordinary skill in the art that the XO signal willalso be asserted when CLK_OUT0 and CLK_OUT1 are 180 degrees out ofphase. To avoid a false detection of crossover, we need to furtherqualify the XO signal using AND gate 1050 and a logical signal OS_GT_TH,which is asserted only when the absolute value of the phase offsetvariable PHA_OS is greater than a predetermined threshold PHA_TH. ABS1080 and CMP 1090 are used to generate the logical signal OS_GT_TH,which is indicative of whether or not the absolute value of PHA_OSexceeds the threshold value PHA_TH.

In the embodiment illustrated in FIG. 9, we use a crossover detector todetermine a crossover condition, upon which we must assert the logicalsignal RESET and toggle the STATE signal. In an alternative embodimentwithout using an explicit crossover detector circuit, we assert thelogical signal RESET when the phase offset variable PHA_OS within FSM710 corresponds to a phase offset of 180 degrees. For example, we expecta crossover condition to occur when the value of PHA_OS·J/I equals ½ or−½, where J is a magnitude of current for each ternary DAC cell withinDAC 920 (of FIG. 9) and I is a magnitude of charge pump current withinPD 310_0 and PD 310_1. In this alternative embodiment, we predict acrossover condition in an open-loop manner. The prediction will be veryaccurate if the matching of current magnitude among the charge pumpcircuits within DAC 920 and the charge pump circuits within PD 310_0 andPD 310_0 is good.

In a further embodiment, the inverted input clock CLK_INB is not exactly180 degrees out of phase relative to the input clock CLK_IN. Forexample, it can only be 90 degrees out of phase relative to the inputclock CLK_IN. The method disclosed and illustrated in FIG. 9 will stillwork as long as the crossover condition is properly detected.

For those of ordinary skill in the art, the principle disclosed by thepresent invention can be practiced in various forms. For example, onemay employ three DLCS: one of them is in an active state while the othertwo are in a stand-by state, and exchange the roles of the active DLCSand one of the two stand-by DLCS when a crossover condition is detected.Using more than two DLCS, however, increases hardware complexity whileoffering no significant advantages and therefore is not attractive.Also, a DLCS is just an example of a variable delay clock circuit. Onecan freely replace DLCS 300_0 (or DLCS 300_1) by any variable delayclock circuit, as long as the variable delay clock circuit receives aninput clock (CLK_IN or CLK_INB) and an offset signal (PO0 or PO1) andgenerates an output clock (CLK_OUT0 or CLK_OUT1) that has a phase offset(relative to its input clock, CLK_IN or CLK_INB) determined by theoffset signal (PO0 or PO1).

Those skilled in the art will readily observe that numerousmodifications and alterations of the device and method may be made whileretaining the teachings of the invention. Accordingly, the abovedisclosure should be construed as limited only by the metes and boundsof the appended claims.

1. An apparatus comprising: an adjustable delay circuit for receiving aninput clock and for generating an output clock having a phase offsetdetermined by a control signal; a phase detector for detecting a phasedifference between the input clock and the output clock and forgenerating a phase error signal representing the phase difference; anencoder for receiving an input code and thereby converting the inputcode into a plurality of output codes; a plurality of converters forconverting said output codes into a plurality of analog signals,respectively; a summing circuit for summing the phase error signal andthe plurality of analog signals into a modified phase error signal; anda filter for filtering the modified phase error signal to generate thecontrol signal.
 2. The apparatus of claim 1, wherein at least one ofsaid output codes is a binary code.
 3. The apparatus of claim 1, whereinat least one of said output codes is a ternary code.
 4. The apparatus ofclaim 1, wherein at least one of said converters is a charge pumpcircuit.
 5. The apparatus of claim 1, wherein the phase error signal hasa sign indicative of a timing relationship between the input clock andthe output clock and has a magnitude substantially proportional to thephase difference between the input clock and the output clock.
 6. Theapparatus of claim 1, wherein the phase detector is a linear phasedetector.
 7. The apparatus of claim 1, wherein the adjustable delaycircuit is a voltage controlled delay line.
 8. The apparatus of claim 1,wherein the phase detector further comprises a charge pump circuit andthe phase error signal is in a form of an electrical current.
 9. Theapparatus of claim 1, wherein at least one of the plurality of analogsignals is in a form of an electrical current.
 10. A method ofgenerating an output clock, the method comprising: receiving an inputclock; generating the output clock by delaying the input clock by anamount of delay controlled by a control signal; detecting a phasedifference between the input clock and the output clock to generate aphase error signal; converting an input code into a plurality of outputcodes; performing a plurality of digital-to-analog conversions toconvert said output codes into a plurality of analog signals,respectively; summing the phase error signal and the plurality of analogsignals into a modified phase error signal; and filtering the modifiedphase error signal to generate the control signal.
 11. The method ofclaim 10, wherein at least one of said output codes is binary code. 12.The method of claim 10, wherein at least one of said analog signals isin a form of an electrical current.
 13. The method of claim 10, whereinthe phase error signal has a sign indicative of a timing relationshipbetween the input clock and the output clock and a magnitudesubstantially proportional to the phase difference between the inputclock and the output clock.
 14. The method of claim 10, wherein thecontrol signal is in a form of a voltage.
 15. The method of claim 10,wherein the phase error signal is in a form of an electrical current.16. The method of claim 10, wherein at least one of the plurality ofanalog signals is in a form of an electrical current.
 17. The method ofclaim 10, wherein at least one of said output codes is ternary code.